1. Introduction

From the rapid development of wireless communication many new standards emerge operating at different frequency bands. IMT-advanced (international mobile communications advanced), which is the 4th generation mobile communication system, has been the focus of the research into new generation wireless communication. UWB, known for its low power, high data rate and lower cost, is a promising technology for wireless communication and local area networks^[1]. With the recent advances in CMOS technologies, the possibility of integrating a monolithic chipset with multi-standards and multi-bands transceivers has been increased as a consequence of the rapid growth of modern wireless communication.

The carrier frequencies increase from several gigahertz to over ten gigahertz. As the frequency increases, the synthesizer is required to work in a wide band with good phase noise. The VCO part of the synthesizer dominates the block's phase noise performance. In a typical differential LC VCO, two main noise sources contributed to the phase noise. One is the tail current, and the other is the negative $g_{\rm m}$-cell^[2]. Some studies have revealed that the tail current source noise is an important contributor to the phase noise, whereas the 1/$f$ noise of the $g_{\rm m}$-cell does not account for the close-in phase noise^[3]. To improve the VCO phase noise, a number of methods have been proposed. Complementary cross-coupled topology is used for better phase noise due to its more symmetric oscillation waveforms^[4]. Removing the bias current source also helps to improve phase noise, while it has a problem of increasing the sensitivity to power supply noise. The reduced minimum device length of the CMOS process results in a lower supply voltage, which limits the tuning control voltage and hence the achievable tuning range. In order to extend the LC VCO tuning range, many researchers have presented a variety of techniques. Though multiple VCOs can be used to generate multiple frequency bands, the most common technique is utilizing a switched capacitor array due to a smaller chip size and lower cost. The target frequency tuning range can be split into several sub-bands, which can decrease the VCO tuning gain and lower phase noise. Moreover, shorter length devices exhibit lower trans-conductance gain, which involves a higher bias current to fulfill the start-up conditions. Generally, one accepts a degraded noise performance, in exchange for ensuring operation over a larger tuning range.

Based on the TSMC 0.13 $\mu$m CMOS process, this paper presents a low phase noise VCO for IMT-advanced (3.4-3.6 GHz) and UWB (4.2-4.8 GHz) systems. The proposed VCO was implemented using a switched current source, noise filter and triode-region MOSFET techniques, which are effective to reduce the power dissipation with the minimum degradation of area efficiency and phase noise. The reconfigurable LC tank has been used to simply adjust the oscillation frequency of the VCO by the switched capacitors. For dual-band applications, a switched current source has been used to guarantee a reliable startup. The phase noise improvement technique is applied to a tail current bias source and cross-coupled differential pair. The measured result exhibits a good phase noise of -123 dBc/Hz at 1 MHz offset for IMT band and -119 dBc/Hz at 1 MHz offset for the UWB band.

2. Design of the LC-VCO

2.1 Circuit architecture

Figure 1 is the topology of the conventional LC-VCO. The complementary cross-coupled topology shown in Fig. 1(a) offers a better phase noise performance for a given bias current^[5]. However, the complementary cross-coupled topology requires three tiers of MOSFET, it is not suitable for a low power supply due to limited voltage redundancy^[6]. Another drawback of complementary cross-coupled topology is the extra $V_{\rm GS}$ needed to operate.

The prime design considerations for the proposed VCO aimed at improving the tuning range and area efficiency, as well as minimizing phase noise. To implement a wide tuning range of the LC-VCO, it is critical to decrease the parasitic capacitors. Considering the same $g_{\rm m}$, the width to length ratio ($W/L$) of the NMOS cross-couple is about one third of that of the PMOS cross-coupe. Thus, the parasitic capacitor in a NMOS pair is less than that in a PMOS pair due to the smaller transistor size.

In this design, an all NMOS cross-coupled LC oscillator is chosen due to the fact that the parasitic capacitor is lower than the PMOS transistor. By careful design, an all NMOS cross-coupled LC oscillator can achieve a balanced tuning range and phase noise performance for a low voltage supply. The proposed VCO topology is shown in Fig. 1(b). A four bits switching capacitors array is introduced to reduce the gain of voltage control and meanwhile to optimize the phase noise performance.

2.2 LC tank design

The oscillation frequency of the LC-VCO is determined by the LC tank, as in Eq. (1). Here $C_{\rm V}$ is the variable capacitor, and $C_{\rm P}$ is the parasitic capacitor. In the CMOS process, the $Q$ factor of the LC tank is limited by the chip inductor^[7]. The differential inductor has been used here since its $Q$ factor is higher.

$f_{\rm OSC} \approx \frac{1}{2\pi \sqrt {L_{\rm tank} \left( {C_{\rm fix} +C_{\rm v} +C_{\rm P}} \right)}}.$

(1)

Under high frequency working conditions, the LC tank current swings as a sine wave due to the latency of the transistor switching on. Then the oscillator differential waveform is shown in Eq. (2).

$V_0 =IR_{\rm p} =I\omega _0 LQ,$

(2)

where $Q$ is the LC tank quality factor, $R_{\rm p}$ is the LC tank shunt resistance, and $\omega_{0}$ is the oscillation frequency. Considering the power consumption, to increase the oscillation amplitude, $Q$ and $L$ need to be enlarged. Normally, the $Q$ factor is set at maximum, as this is the only way to increase the inductance. As there is a choice of tank inductor, the design strategy is to achieve simultaneous optimization on both the phase noise and tuning range with the constraint of minimum power consumption. The inductance can not be too large since the parasitic capacitor will affect the LC tank^[8]. Initially, the inductance selection scheme assumes the phase noise is the main design goal. Afterwards, the tuning range and power consumption constraints will be considered to set the final tank inductor. This is an iterative design strategy. After simulation, a 60 $\mu$m inner circle radius, with a 15 $\mu$m width, and 2 laps inductor is chosen, with a $Q$ factor of 15.

The varactor geometrical parameters also need to be accurately chosen in order to optimize the quality factor $Q_{\rm VAR}$. Indeed, at high operating frequencies, $Q_{\rm VAR}$ could significantly decrease the overall tank quality factor^[9]. For this purpose, minimal-length and minimal-width varactors have been employed.

To satisfy a sufficient tuning range, the only tuning varactor cannot fulfill the requirement, and the high gain varactor will degrade phase noise performance^[10]. Therefore, a four bits switching capacitor array is used to extend the tuning range, as shown in Fig. 2. The capacitance switches, SW$_{\rm C1}$-SW$_{\rm C3}$ are for extending a given frequency band; to accomplish dual-band operation, the capacitance switch, SW$_{\rm CB}$, is for band selection. An MIM capacitor is used in the designed VCO as the fix capacitor due to its high $Q$ factor.

2.3 Switched current source and cross couple pair design

For an LC VCO, the start-up condition is given by the well known inequality $g_{\rm m}$ $\geqslant$ 2/$R_{\rm p}$. Ignoring the parasitic resistor and capacitor, the LC tank resistor $R_{\rm p}$ is equal to $Q\omega L$, where $R_{\rm p}$ is a frequency dependent factor that cannot be ignored in the wideband oscillator design. At the low end of the IMT-Advanced band, the $R_{\rm p}$ reaches its lowest value^[11], thus more current is required from cross coupled pair transistors in order to increase the transconductance. The switching capacitor array worsens the oscillation tank $Q$ factor, which decreases the LC tank resistor $R_{\rm p}$ as well.

The switch current source is widely used in wideband VCO design in order to guarantee the oscillation over the whole band. A traditional switch current source biased VCO is shown in Fig. 3(a). The switch current source will be switched on to compensate more energy loss in the LC tank as the switch capacitor increases. This compensation will keep the VCO oscillate at a low band and a stable oscillation amplitude.

It is well known that flick noise from the tail current source will be the main contributor to 1$/f^{3}$ phase noise. Flick noise is correlated noise and can only exist in systems with memory. When a cross couple pair are ideally switched, all memory and consequently the flicker noise is removed^[13]. In Fig. 3(a) design, there is a limitation that as the cross couple current increases, the overdriving voltage of the transistor pair of MC1 and MC2 will increase as well, this voltage increasing will force a current source into the triode region. Current source in the linear region affects the cross pair switching characteristic thus flicker noise needs to be considered. If the cross pair transistors switching is not ideal, a small amount of NMOS transistor flicker noise will be upconverted.

In this design, a switched cross coupled NMOS pair has been used to guarantee the worst case start-up conditions, and this NMOS pair is controlled by the switched current source. As shown in Fig. 3(b), the current source of $I_{\rm cst}$ is on over the whole frequency tuning range, MC1 and MC2 compensate the energy loss of the LC tank. At IMT-A band, a switch signal forces the extra current source of $I_{\rm SW1}$ on, a cross couple of MC3 and MC4 will provide more energy to the LC tank, and this enlarges the total $g_{\rm m}$.

The main advantage of this scheme is that the switched current source could not affect the DC operation point of the current source $I_{\rm cst}$. Thus, the switch current source can work in the saturation region over the whole frequency band, then the cross couple pair which provides the negative resistance can also work in a good switching mode, this will reduce the flick noise effect from the tail current source. Another advantage is that the switch controlled cross-coupled NMOS pair only works at a low frequency band, so this will not add extra power consumption when the VCO works at a high frequency band.

Compared with a traditional one, the switch cross coupler current source brings in a larger parasitic capacitor, so the size of the transistor pair is limited by the VCO tuning range. The width to length ratio of the cross coupler and switch current source can be selected by the below principle: the oscillation amplitude of the VCO is equal at the highest and lowest frequency; and the common mode voltage of A, B shown in Fig. 3(b) is equal.

Using the minimum size of transistors to help improve the phase noise performance and meanwhile decrease the parasitic capacitor. Under some conditions, the size and area of the cross coupler pair can be larger if the $1/f$ noise is too high^[13]. Simulation results show that the $1/f$ noise hardly affects the VCO phase noise, so the minimum length of the cross coupler pair is selected due to the consideration of reducing the parasitic capacitor and decreasing the power consumption.

2.4 Phase noise optimization

In IMT-Advanced, signal EVM (error vector magnitude) is a key factor. Phase noise is the key performance of a VCO, it has a small impact on the signal EVM. As shown in Fig. 4, two inductors ($L_{1}$, $L_{2})$ are placed at nodes A and B, respectively. Inductors are chosen to resonate with the MIM capacitor ($C_{1}$ and $C_{2})$ and parasitic capacitance at frequency of 9GHz and 7 GHz, which are the second harmonic frequency at the centers of the two working bands, so the impedance at the node is high at frequency 2$\omega_{0}$. The reason for choosing 2$\omega_{0}$ as the resonation frequency can be explained as follows. Flicker and thermal noise from the tail current source which enters the LC tank will be upconverted due to the mixing action of the VCO circuits. Also, if the differential oscillator is unbalanced, the common-mode node of the current source will oscillate at twice the oscillator center frequency, because the current source will be pulled every time one of the NMOS cross couple pair is switched on. This 2$\omega_{0}$ noise enters the LC tank and mixes with the fundamental oscillator frequency, and results in a phase noise side band at the oscillator frequency. Therefore, to minimize the phase noise introduced by upconversion, all even harmonics must be suppressed. This noise filter rejects thermal noise in the tail current source around the second harmonic of the center frequency^[14]. Then the performance of phase noise is improved.

Additionally, bias filtering is also used in the scheme to suppress the noise from reference current noise and flick noise on the VCO current source. It can also suppress the dynamic transient effects on the bias line. The passive components resistor and capacitor $R_{\rm f}$, $C_{\rm f}$ form a low-pass filter to suppress noise and transients from the reference current.

The full schematic of the proposed VCO is shown in Fig. 4. The LC tank is composed of a center tapped inductor, a switched capacitor array, as shown in Fig. 2 and two accumulation mode MOS varactors ($C_{\rm V})$. Cross-coupled NMOS and switched cross-coupled NMOS pairs in the VCO provide the transconductance to compensate for the LC tank losses. The noise filter consists of two standard inductors ($L_{1}$, $L_{2})$, two MIM capacitors ($C_{1}$, $C_{2})$, and a poly resistor ($R$).

3. Measurement results

The proposed wide band VCO is fabricated in a TSMC 0.13 $\mu$m CMOS process. The microscope photograph of the designed VCO is shown in Fig. 5. The chip size is 1.11 $\times$ 0.98mm$^{2}$ with pad included. A chip test with three sets of probes is used to verify chip performance.

The oscillation waveforms of the designed VCO at two bands are shown in Fig. 6. The oscillation waveform peak to peak amplitude are 886.4 mV and 932.8 mV, this indicates that the VCO can work properly at two bands with little amplitude difference. Supplied by a 1.2 V DC voltage, the designed VCO consumes 5.6 mA and 3 mA at the IMT-A and UWB bands respectively.

The tuning characteristics of the designed VCO at the UWB band and the IMT-Advanced band are tested. The discrete tuning characteristics of each frequency band by a combination of the capacitor switches are overlapped. The low frequency band and the high frequency band can be tuned to 3.14-3.88 GHz and 3.86-5.28 GHz continuously. Compared with the simulation result, there is a 200 MHz frequency range decrease. The root cause of this is inductor modeling, it is inaccurate and the parasitic inductor can not be extracted in the simulation. Since enough of a tuning range margin is reserved during the VCO simulation, the desired working band can still be covered by this designed VCO.

The phase noise performance of the designed VCO is shown in Fig. 7, the measured phase noise of 1 MHz frequency offset are -122.06 dBc/Hz and -119 dBc/Hz at the IMT-Advanced and UWB bands respectively. The phase noise performance is not as good as that in simulation. This may caused by the fact that the $Q$ factor of the inductor is worse than in the simulation, the varactor modeling is not accurate enough, and there is additional noise which was brought in by the power supply and can be modulated into phase noise.

The figure-of-merit (FoM) in Table 1 is commonly used to compare the VCO with others, it suits the 1/$f^{2}$ region phase noise comparison^[15], which is defined as:

${\rm{FoM}}(\Delta \omega ) = L(\Delta \omega ) - 20\lg \frac{{{\omega _{\rm{o}}}}}{{\Delta \omega }} + 10\lg {P_{{\rm{mW}}}}.$

(3)

In this formula, $\omega_{0}$ is the VCO center frequency, $L(Δ \omega)$ is the phase noise at $Δ \omega$ offset, and $P_{\rm mW}$ is the VCO power consumption. The higher the FOM, the better the VCO performance.

4. Conclusion

In this paper, a wideband LC VCO is designed in a TSMC 0.13 $\mu$m CMOS process. For IMT-Advanced and UWB band applications, a switch current source has been used to fulfill the start-up condition in a low band. Bias filters are used in the VCO to suppress the reference current noise and dynamic transient effects on the bias line. The proposed VCO performs a phase noise of around -120 dBc/Hz at 1.0 MHz offset frequency over the whole working band, and the tuning range of 3.14-3.88 GHz and 3.86-5.28 GHz. The current consumption is less than 4.5 mA from the 1.2 V power supply. The designed VCO works well in the required band with performance satisfying system requirements.